A Color LED Driver Implemented by the Active Clamp Forward Converter

Abstract

Because light emitting diodes (LEDs) have the advantages of dc working voltage, high luminescent efficiency, short ignition time, high reliability and pollution free, they have substituted for incandescent bulbs and fluorescent lamps gradually. In order to simplify circuit complexity, an active clamp forward converter with the sequential color display (SCD) control is proposed to drive red, green and blue (RGB) LED arrays. The proposed converter has zero-voltage switching (ZVS) operations of both the main switch and the auxiliary switch, resulting in high system efficiency. Driving RGB LED arrays sequentially by one converter can save components and reduce cost significantly. Additionally, the pulse-width modulation (PWM) control is applied to achieve a large chromaticity variation. The circuit operations are analyzed in detail and the circuit parameters are designed based on the practical considerations. Finally, an illustrative example is implemented to demonstrate the feasibility and validity of the proposed LED driver.

1

Introduction

Recently, light emitting diodes (LEDs) have substituted for incandescent bulbs and fluorescent lamps gradually in the residential, industrial and commercial lighting applications [1], because they have the advantages of dc working voltage, high luminescent efficiency, short ignition time, high reliability and pollution free, etc. Moreover, since mixing multi-coloured LEDs can provide a large chromaticity variation, LEDs are progressively used in outdoor landscape lighting systems as well.

Figure 1 shows the block diagram of a photovoltaic powered landscape lighting system, in which an isolated DC/DC converter is used to fulfill a step-down conversion, govern the current through LED arrays and provide the electrical isolation of safety requirements. Generally, a colored LED module is composed of red, green and blue (RGB) LED arrays. Since RGB LEDs have different electrical characteristics, three isolated DC/DC converters are required to regulate the current through each LED arrays [2]. Therefore, the number of power components will become triple, resulting in large volume, high cost and low reliability.

Figure 1.

The block diagram of a photovoltaic powered landscape lighting system.

In the literature [3, 4], a sequential color scanning scheme was proposed to reduce power consumption. LEDs are very suitable for a sequential color display (SCD) due to their fast response. When LEDs are displayed intermittently with high frequency, human cannot observe the darkness of short time intervals. The SCD control has been applied in a color LED backlight driving system for liquid crystal display (LCD) panels [5], [6], in which RGB LED arrays are sequentially driven by the same DC/DC converter.

Many isolated DC/DC converters for RGB LED driving, such as flyback and forward converters, have been presented in the literature [3, 5].However, their hard-switching operations reduce conversion efficiency and induce electromagnetic interference (EMI) noise. This paper proposes an active clamp forward converter [7-9] with SCD control as the color LED driver for a photovoltaic powered landscape lighting system. For this driver, due to saving transformers, power switches, control ICs and energy storage components, the circuit complexity can be simplified, and the system volume and cost can be significantly reduced. With accurate analysis and design, both of the main switch and the auxiliary switch can turn on under zero-voltage switching (ZVS), leading to higher system efficiency and lower EMI noise. Besides, the pulse-width modulation (PWM) control is applied to change the luminance of RGB LEDs so that a large chromaticity variation can be achieved. The operation principles are addressed. Experimental results of a 100 W laboratory prototype are used to verify the feasibility and validity of the theoretical predictions.

2

Operation principles

The proposed color LED driver, as shown in Figure 2, consists of an active clamp forward converter and RGB LED arrays with SCD control. In this section, the operation principles of the SCD control and the proposed converter are introduced briefly in the following.

Figure 2.

The conceptual circuit of the proposed color LED driver with SCD control.

2.1

Sequential color display control

To achieve SCD control, RGB LED arrays are connected in series with the switches S_R, S_G and S_B, respectively. The switches are driven with phase-shift manner, as illustrated in Figure 3, in which v_gs__r, v_gs__gand v_gs__bare their gate signals with the switching frequency f_m. The frequency of the driving signals v_GS1 and v_GS2 in the active clamp forward converter is f_s which should be much higher than f_s. The voltages V_{o_R}, V_{o_G} and V_{o_B} are the forward voltage of RGB LED arrays, respectively. The currents i_R, i_G and i_B are their corresponding driving currents.

Figure 3.

Driving signals and conceptual key waveforms of the SCD control.

Assuming that the currents i_R, i_G and i_B have the same amplitude l_P at full-load condition, the maximum average output-current l_o,avg of a conventional RGB LED driver is 3l_P. For the proposed driver, if the maximum duty cycle of the driving signals v_GS__R, v_GS__G and v_GS__B are D_max, then the maximum average output-current l_o,avg is

Because D_max is 33.33% in the proposed driver, l_o,avg is only equal to l_P. Therefore, the required output-power of the proposed driver is only one-third of that of a conventional driver, which results in energy saving significantly.

In the proposed system, the red, green and blue LED arrays are driven by the same converter sequentially, leading to small volume, low cost and high reliability. All of the driving signals v_gs__r, v_gs__gand v_gs__bare produced by the digital signal processor (dsPIC30F4011). The luminance of each LED array is dimmable by modulating the duty of its driving signal, so as the color of the RGB-LED module is variable. The output capacitor C₀ with low capacitance is used in the converter so that the output voltage can vary promptly according to the forward voltage of each LED array.

2.2

Active clamp forward converter

Figure 4 shows the circuit diagram of the active clamp forward converter. Each active switch S₁ (S₂) is composed of an MOSFET Q₁ (Q₂) and its intrinsic anti-parallel diode D₁ (D₂). The forward converter is formed by the main switch S₁, transformer T_x, forward diode D₃, freewheel diode D₄, output inductor L₀, and output capacitor C_o. Instead of the reset winding in traditional forward converter, the auxiliary switch S₂ and the clamping capacitor C_c are used to reset the residual flux of transformer. The residual energy of magnetizing inductor L_m can be restored to the input voltage source. Both of the main switch S₁ and the auxiliary switch S₂ can turn on under ZVS, resulting in low switching losses and high system efficiency.

Figure 4.

Circuit diagram of the active clamp forward converter.

For simplifying the analysis of operation principles, following conditions are made:

• 1)

  All components are ideal.

• 2)

  The capacitance ofC_c is much higher than that of the intrinsic capacitorC_ds, and the inductance ofL_m is much higher than that of the leakage inductorL_r.

• 3)

  L_o and C_o are large enough so that the output voltageV_o and output currentl_o can be regarded as constant.

• 4)

  The dead time between the driving signals ofS₁ andS₂ is very short and negligible. The turn-on intervals ofS₁ andS₂ are considering asDT_S and (1-D)T_S, in whichT_s is the switching period in steady-state operation.

According to the volt-second balance concept, the relation of input voltage V_B and the voltage across clamping capacitor v_c can be expressed as

Arranging Eq. 2 yields

Eq. 3 indicates that v_c is the output of boost-type operation with input voltage V_B. When S₁ turns off and S₂ turns on, the voltage across main switch v_ds1 will be equal to v_c and determined as

Figure 5 shows the theoretical waveforms of the active clamp forward converter, which is divided into nine operation modes within one switching period. The operation principle of each mode will be introduced referring to the equivalent circuits shown in Figure 6.

Figure 5.

Theoretical waveforms of the active clamp forward converter.

Figure 6.

Equivalent circuits of operation modes.

Mode 1 (t₀ ~ t₁): At the moment t = t₀, the switch S₁ is turning on, and the switch S₂ is turning off. Since the voltage across magnetizing inductor v_n1 is equal to V_B, its current i_Lm increases linearly. The voltage across secondary winding v_n2 is V_BI n so that the forward diode D₃ turns on and the freewheel diode D₄ turns off. The input voltage source charges output inductor L_o through the transformer T_x and D₃. This mode ends while the gate signal of S₁ disappears.

Mode 2 (t₁ ~ t₂): At the moment t = t₁, the main switch S₁ turns off, and the current i_Lr charges the intrinsic capacitor C_ds of main switch. The voltage v_ds1 increases, and the voltage v_n1 decreases. This mode ends while v_ds1 reaches V_B.

Mode 3 (t₂ ~ t₃): At the moment t = t₂, the voltage v_n1 reaches zero so that D₃ and D₄ are both turn-on. Since the diode current i_D3 decreases, the leakage inductor current ¡_Lr decreases as well. The magnetizing inductor current i_Lm remains the same. This mode ends while v_ds1 reaches V_BI (1-D).

Mode 4 (t₃ ~ t₄): At the moment t = t₃, the main switch S₁ turns off. The current i_Lr flows through anti-paralleled diode D₂ to transfer energy to the clamping capacitor C_c, which provides ZVS operation for the auxiliary switch S₂ turn-on. Gate signal of S₂ should be applied during this mode. This mode ends while i_D3 reaches zero.

Mode 5 (t₄ ~ t₅): At the moment t = t₄, because the current i_D3 is zero, the inductor L_m is in series with L_r and participates in the resonance with C_c. The voltage v_n1 is DV_BI (1-D) with reverse polarity. The resonant current i_Lr is equal to the current i_Lm and remains decreasing. While i_Lr becomes negative during this mode, the energy of L_m will be restored to the input voltage source. This mode ends while the gate signal of S₂ disappears.

Mode 6 (t₅ ~ t₆): At the moment t = t₅, the auxiliary switch S₂ turns off, and the resonant current i_Lr follows through the intrinsic capacitor C_ds to discharge it. This mode ends while the voltage v_ds1 decreases to V_B.

Mode 7 (t₆ ~ t₇): At the moment t = t₆, the voltage v_ds1 is equal to V_B so that the voltage v_n1 becomes zero again. Since D₃ are turning on now, its current i_D3 increases and the current i_Lr increases as well. The magnetizing inductor current i_Lm remains the same. This mode ends while v_ds1 reaches zero.

Mode 8 (t₇ ~ t₈): At the moment t = t₇, the current i_Lr-flows through anti-paralleled diode D₁ to transfer energy back to the input voltage source, which provides ZVS operation for the main switch S₁ turn-on. Gate signal of S₁ should be applied during this mode. The current i_Lr increases from negative with the slope of V_BI L_r. This mode ends while i_Lr- reaches zero.

Mode 9 (t₈ ~ t₉): At the moment t = t₈, the current i_Lr becomes positive. The current i_D4 is decreasing continuously, and this mode ends when i_D4 reaches zero. When the diode D₄ is turned off at t = t₉, the operation goes to Mode 1 of the next high-frequency switching cycle.

3

Design procedures

Based on the operation principles presented previously, the proposed color LED driver with the specifications listed in Table 1 is taken as an illustrative example. The designed procedures of the active clamp forward converter are described as follows.

Table 1.

Electrical Specifications of the Prototype.

input voltage, Vb	18 ~ 36 Vdc
output voltage, Vo	11.4 Vdc
maximum output current, lo	2 A
ripple of output voltage, ΔVo	10 mVpp
switching frequency, fs	150 kHz

Step 1-Select f_s and D: Select the switching frequency f_s as 150 kHz. For the normal operation condition with 24 V battery voltage and full load output, the duty ratio D of the main switch S₁ is determined as 0.5.

Step 2-Determine n: Considering that the forward voltage V_f of the diode D₃ is around 0.7 V, the minimum turn ratio (n) of the transformer T_x can be determined as

In the illustrative case, (n) is set as 1.

Step 3-Determine L_o: In order to minimize the output ripple current and reduce the current stress of power components, we select 10% of maximum output current as the boundary condition. The boundary inductance of L₀ can be obtained as

In the illustrative case, L_o is set at 100 μH.

Step 4-Determine C_o: While output inductance L_o is determined, considering that the specification of ΔV₀ is 10 mV, the capacitance of C_o can be obtained

The standard capacitance with 33 μF is selected for C_o.

Step 5-Determine n₁: Select core material as TDK PC40 and core size as EE-19. Its saturation flux density B_sat is 4500 G at 60 °C, and effective cross area A_e is 0.23 cm². To avoid core saturation, we set the maximum flux density B_max at half of B_sat. Therefore, the turn number of primary winding n₁ can be calculated as

In the illustrative case, n1 is set as 8. The inductance of L_m can be measured as 80 μ.

Step 6-Determine L_r: According to the operation principle of Mode 6, the current i_Lr at t = t₆ can be obtained from following

To achieve ZVS operation of the main switch S₁, the inductance of L_r should meet following equation

Considering that the intrinsic capacitor C_ds of IRF530 is 190 nF, the required inductance of L_r can be obtained as 438 nH.

Step 7-Determine C_c: To optimize the voltage stresses of C_c, S₁ and S₂, the period of the resonance occurring during S₁ turning-off should be longer than ten times of the turn-off time of S₁, which is expressed as

Thus, the minimum capacitance of C_c can be obtained as

The standard capacitance with 470 nF is selected for C_c.

The calculated component parameters of the illustrative example are summarized in Table 2.

Table 2.

Component Parameters of the Prototype.

main switch, S1	IRF530(100V/10A)
auxiliary switch, S2	IRF9540(100V/19A)
diodes, D3, D4	SB1060(60V/10A)
leakage inductor, Lr	438 nH
magnetizing inductor, Lm	80 μ
clamping capacitor, Cc	0.47 μF
output inductor, Lo	100 μ
output capacitor, Co	33 μF
transformer turn ratio, n:1	1:1

4

Experimental results

While input voltage V_B is 24 V, Figure 7 shows the measured waveforms of the output voltage V_o and the output inductor current i_Lo at full load operation. It can be seen that V_o is regulated at 11.41 V, and the average output current is 2.04 A, which are very close to the required specifications. The measured waveform of clamping capacitor voltage V_c according the driving signal V_GS1 is shown in Figure 8. It is obvious that resonance with the frequency determined by C_c and (L_r + L_m) occurs after the main switch S₁ is turning off. In this time interval, the resonant current i_Lr transfers energy from L_m to C_c, and then restores energy to the input voltage source.

Figure 7.

Measured waveforms of the output voltage V_o and the output inductor current i_Lo at full load operation (V_o. 5 V/div; i_Lo: 1 A/div; time: 2 μs/div).

Figure 8.

Measured waveforms of clamping capacitor voltage V_c according the driving signal V_GS1(V_gsï:10 V/div; V_c. 20 V/div; time: 1 μs/div).

Figure 9 shows the measured voltage and current waveforms of the main switches S₁ at full load condition. It could be observed that S₁ can turn on under ZVS condition. Figure 10 shows the measured voltage waveforms of the auxiliary switches S₂ at full load condition. The drain to source voltage v_ds2 becomes zero before the driving signal V_GS2 is supplied, from which the ZVS operation of S₂ can be verified. Therefore, the switching losses can be significantly minimized to improve system efficiency.

Figure 9.

Measured voltage and current waveforms of the main switch S₁ at full load operation (V_ds1: 50 V/div; i_s1: 5 A/div; time: 1 μs/div).

Figure 10.

Measured voltage waveforms of the auxiliary switch S₂ at full load operation (V_ds2: 20 V/div; V_gs2: 5 V/div; time: 1 μs/div).

Figure 11 shows the measured efficiency curve of the proposed color LED driver with 24 V input voltage. The maximum efficiency is up to 92.5% at /₀ = 1.6 A. The normal operating range for driving color LED arrays is from 1.2 A to 1.8 A. The system efficiency can be kept above 91% in this range, which provide energy saving significantly.

Figure 11.

Measured efficiency curve of the proposed color LED driver.

Figure 12 shows the measured waveforms of output voltage V_o and output current I_o while the LED module outputs “white” light. It can be seen that red, green, and blue LED arrays are driving in sequential and with the frequency of 30 Hz. The output voltage V_o is around 10.4 V for driving red LEDs, and V_o is regulated to around 11.4 V for driving green and blue LEDs. The duty cycle of each LED is almost 100% so that all of them provide the highest luminance to have “white” light output. In Figure 13, the duty cycle of green LEDs is changed to be 50% to reduce the luminance. Since the luminance of red and blue LEDs remain 100%, the color of the light provided by LED modules is “purple”. These experimental results verified that a large chromaticity variation can be achieved by low-frequency PWM control. By the way, the photo of the illustrative prototype is shown in Figure 14.

Figure 12.

Measured waveforms of output voltage V_o and output current l_o while the LED module outputs “white” light (V_o: 2 V/div; l_o. 500 mA/div; time: 10 ms/div).

Figure 13.

Measured waveforms of output voltage V_o and output current l_o while the LED module outputs “purple” light (V_o: 2 V/div; /_o: 500 mA/div; time: 10 ms/div).

Figure 14.

Illustrative prototype of the proposed color LED driver.

5

Conclusion

An active clamp forward converter with the SCD control for driving RGB LED arrays has been proposed. ZVS turn-on operations of both the main switch and the auxiliary switch lead to low switching losses and high system conversion efficiency. RGB LED arrays are sequentially driven by the same converter, which can save components and reduce cost significantly. Besides, PWM control is applied to achieve a large chromaticity variation. Experimental results of a 100 W illustrative example have really proved the feasibility and validity of the theoretical predictions.

Acknowledgements

This work was supported by the National Science Council of Taiwan, Republic of China, under grant contract NSC 101-2221-E-214-059.